Electrical power conversion apparatus

ABSTRACT

An electrical conversion apparatus including a converter and an inverter comprises a capacitor that stores a DC electrical power; a ripple detection unit that detects a ripple of an RMS power that is generated from the inverter; a voltage measuring instrument that measures a voltage across the capacitor; a DC voltage command generation unit that calculates a command value of the voltage across the capacitor according to a frequency of the AC voltage generated from the inverter; and a DC voltage control unit that receives the voltage measured by the voltage measuring instrument and the command value calculated by the DC voltage command generation unit, to control the converter so that the voltage across the capacitor becomes the command value, wherein the DC voltage command generation unit makes the command value of the voltage across the capacitor higher than usual, in situations where the voltage across the capacitor is within a predetermined range including a frequency of a ripple component of the voltage across the capacitor.

TECHNICAL FIELD

The present invention relates to electrical conversion apparatuses thatconvert a direct current (DC) electric power to a variable-frequency,variable-voltage alternating current (AC) electric power, and inparticular to an AC-AC electrical power conversion apparatus thatincludes a converter and an inverter that converts a DC output electricpower generated from the converter into a variable-frequency,variable-voltage AC electric power.

BACKGROUND ART

A PWM converter for use in an electric railway vehicle receives an ACinput from a single-phase AC power source between an overhead line and arail by way of a pantagraph, a transformer and the like, to convert thepower from the source into power of a predetermined DC voltage. Acapacitor for smoothing the voltage is provided on the DC side of thePWM converter, and an inverter for driving an induction motor isconnected to the capacitor. The voltage across the capacitor is detectedby a voltage detector, thereby sensing a DC input voltage that is to beapplied to the inverter. A current detector is provided on the AC sideof the inverter.

A reference output frequency of the inverter is created by addingtogether, using an adder, a rotation frequency—which is an output ofrotation frequency detecting means of the induction motor—and areference slip frequency—which is an output of slip frequency control.An output current value detected by the current detector is provided tocurrent root-mean-square (RMS) value calculating means, therebycalculating a current RMS value. The current RMS value is transmitted tothe adder together with a command current value, thus calculating thereference slip frequency using frequency control means.

A DC input voltage to the inverter is detected by a voltage detector andonly its ripple component is derived from voltage ripple componentdetecting means. The DC input voltage to the inverter is also input toDC voltage component detecting means and only its DC component isderived therefrom. A divider divides the ripple component by the DCcomponent, to calculate a ripple factor of the DC input voltage, and amultiplier multiples the ripple factor by a reference inverter frequencyto calculate an amount of inverter frequency correction. The inverterfrequency is calculated by adding the amount of reference inverterfrequency to the amount of inverter frequency correction using an adder.This inverter frequency is provided to voltage control means, and a PWMcontrol circuit in turn provides a PWM control signal to the inverter(refer to Patent Document 1, FIG. 1 and its corresponding description).

On the other hand, Non-Patent Document 1 has verified the advantageouseffect of Patent Document 1 through experiments. Further, Non-PatentDocument 1 (FIG. 7) provides description of a ripple characteristic of aDC power source in a PWM converter for use in a railway vehicle. As fora relationship between the capacitance of DC capacitor and a suppressioneffect of a beat phenomenon (Non-Patent Document 1 describes based on abeat rate showing how many times a fluctuation range of an inverteroutput current is greater than that in situations where no beat occurs),Non-Patent Document 1 explains that in order to achieve the suppressioneffect such that the beat rate is 1.2 times or less, the ripple factorof DC voltage (a ratio of a DC ripple magnitude over a DC averagevoltage) is reduced to 10% or less. It is described that the capacitanceof DC capacitor must be determined to be approximately 30 mF or more(3750 μF per motor) per 8 motors (output rating of approx. 3000 kW), forinstance.

[Patent Document 1]

Japanese Patent Publication H07-46918 (FIG. 1)

[Non-Patent Document 1]

K. Nakata, A. Kimura, T. Tanamachi, Y. Tsutsui, and K. Nakamura: No. 845“Beat phenomenon in PWM inverter driven on ripple DC power source,”Proceedings of Annual Meeting, I.E.E. Japan, 1988, pp 1039-1040

DISCLOSURE OF INVENTION Problem that the Invention is to Solve

As described above, in the electrical power conversion apparatusaccording to Patent Document 1, the ripple factor of the DC inputvoltage is calculated by dividing its ripple component by its DCcomponent, and the amount of inverter frequency correction is calculatedby multiplying the ripple factor by the reference inverter frequency,whereby the inverter frequency is adjusted according to the ripple ofthe DC input voltage, thereby enabling the current and torque ripple tobe reduced.

However, a problem with the apparatus of Patent Document 1 is that inorder to achieve an effect of reducing the predetermined beatphenomenon, a constraint is applied such that the capacitance of a DCcapacitor is determined, as described in Non-Patent Document 1, so thatthe ripple of the DC voltage can be reduced. That is, since the beatingphenomenon increases at a frequency of twice the AC power sourcefrequency, the capacitance of the DC capacitor is determined by limitingthe ripple component of DC voltage to 10% or less thereof in thefrequency and therefore a problem is that the capacitance of DCcapacitor is increased because of the particular frequency point wherethe beating phenomenon is maximized.

The present invention is directed to overcome the above problem and anobject thereof is to reduce the capacitance of DC capacitor of anelectrical conversion apparatus, as well as to suppress a motor currentripple on the output side of the electrical conversion apparatus andassociated torque ripple.

Means for Solving the Problem

An electrical power conversion apparatus comprises a converter thatconverts an AC power from an AC power source into a DC power; acapacitor that stores the DC power produced from the converter; aninverter that converts to an AC power the DC power stored in thecapacitor; a voltage control unit that calculates command values for anAC voltage to be generated from the inverter, to control the inverter sothat the command value is produced; a current measuring instrument thatmeasures an AC current generated from the inverter; a ripple detectionunit that receives the AC voltage command values calculated by thevoltage control unit and the AC currents measured by the currentmeasuring instrument, to detect a ripple of an RMS power generated fromthe inverter; a voltage measuring instrument that measures the voltageacross the capacitor; a DC voltage command generation unit thatcalculates a command value of the voltage across the capacitor accordingto a frequency of the AC voltage generated from the inverter; and a DCvoltage control unit that receives the voltage measured by the voltagemeasuring instrument and the command value calculated by the DC voltagecommand generation unit, to control the converter so that the voltageacross the capacitor becomes the command value, wherein the DC voltagecommand generation unit makes the command values of the voltage acrossthe capacitor higher than usual in a situation where the frequency of ACvoltage generated from the inverter is within a predetermined rangeincluding a frequency of a ripple component of the voltage across thecapacitor, and the voltage control unit receives the ripple componentderived from the ripple detection unit, to calculate the command valuesfor the AC voltage generated from the inverter so that the ripplecomponent is reduced.

Advantageous Effects of the Invention

An electrical power conversion apparatus comprises a converter thatconverts an AC power from an AC power source into a DC power; acapacitor that stores the DC power produced from the converter; aninverter that converts to an AC power the DC power stored in thecapacitor; a voltage control unit that calculates command values for anAC voltage to be generated from the inverter, to control the inverter sothat the command value is produced; a current measuring instrument thatmeasures an AC current generated from the inverter; a ripple detectionunit that receives the AC voltage command values calculated by thevoltage control unit and the AC current measured by the currentmeasuring instrument, to detect a ripple of an RMS power generated fromthe inverter; a voltage measuring instrument that measures the voltageacross the capacitor; a DC voltage command generation unit thatcalculates a command value of the voltage across the capacitor accordingto a frequency of the AC voltage generated from the inverter; and a DCvoltage control unit that receives the voltage measured by the voltagemeasuring instrument and the command value calculated by the DC voltagecommand generation unit, to control the converter so that the voltageacross the capacitor becomes the command value, wherein the DC voltagecommand generation unit makes the command values of the voltage acrossthe capacitor higher than usual in a situation where the frequency of ACvoltage generated from the inverter is within a predetermined rangeincluding a frequency of a ripple component of the voltage across thecapacitor, and the voltage control unit receives the ripple componentderived from the ripple detection unit, to calculate the command valuesfor the AC voltage generated from the inverter so that the ripplecomponent is reduced. Therefore, in addition to the fact that the motorcurrent and torque ripple on the output side can easily be reduced bydetecting and controlling an AC side ripple that is desirable to reduce,an advantageous effect is that the capacitance of DC capacitor for theelectrical conversion apparatus can be reduced.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram showing an example of a configuration of anelectrical power conversion apparatus according to Embodiment 1 of thepresent invention;

FIG. 2 is a diagram illustrating a configuration of a ripple detectionunit in the electrical power conversion apparatus according toEmbodiment 1 of the present invention;

FIG. 3 is a diagram illustrating a band-pass filter in the electricalpower conversion apparatus according to Embodiment 1 of the presentinvention;

FIG. 4 is a diagram showing a frequency gain characteristic and phasecharacteristic of an example of a band-pass filter in the electricalpower conversion apparatus according to Embodiment 1 of the presentinvention;

FIG. 5 is a diagram showing a DC current command generation unit in theelectrical power conversion apparatus according to Embodiment 1 of thepresent invention;

FIG. 6 is a graph illustrating an operation of the ripple detection unitand the DC current command generation unit in the electrical powerconversion apparatus according to Embodiment 1 of the present invention;

FIG. 7 is a set of graphs showing an effect of reducing a torque rippleby the electrical power conversion apparatus according to Embodiment 1of the present invention. FIG. 7( a) shows a torque waveform generatedby implementing Embodiment 1, FIG. 7( b) showing a torque waveformgenerated when the reduction of the torque ripple is not controlled;

FIG. 8 is a block diagram showing an example of a configuration of anelectrical power conversion apparatus according to Embodiment 2 of thepresent invention;

FIG. 9 is a diagram illustrating a configuration of a ripple detectionunit in the electrical power conversion apparatus according toEmbodiment 2 of the present invention;

FIG. 10 is a diagram showing a DC current command generation unit in theelectrical power conversion apparatus according to Embodiment 2 of thepresent invention;

FIG. 11 is a block diagram showing an example of a configuration of anelectrical power conversion apparatus according to Embodiment 3 of thepresent invention;

FIG. 12 is a diagram showing a DC current command generation unit in theelectrical power conversion apparatus according to Embodiment 3 of thepresent invention;

FIG. 13 is a block diagram showing an example of a configuration of anelectrical power conversion apparatus according to Embodiment 4 of thepresent invention;

FIG. 14 is a diagram showing a ripple detection unit in the electricalpower conversion apparatus according to Embodiment 4 of the presentinvention;

FIG. 15 is a diagram showing a DC current command generation unit in theelectrical power conversion apparatus according to Embodiment 4 of thepresent invention;

FIG. 16 is a diagram showing a switching device in a single armconstituting an inverter according to Embodiment 4 of the presentinvention; and

FIG. 17 is a graph showing a voltage waveform of the switching device ina single arm constituting the inverter according to Embodiment 4 of thepresent invention.

REFERENCE NUMERALS

-   1 Single-phase AC power source-   2 Converter-   3 Capacitor-   4 Inverter-   5 Induction machine (AC rotating machine)-   6 a Current detection unit-   6 b Current detection unit-   6 c Current detection unit-   7 Voltage control unit-   7A Voltage control unit-   8 Ripple detection unit-   8A Ripple detection unit-   9 a Multiplier-   9 b Multiplier-   9 c Multiplier-   10 Adder-   11 RMS power calculation unit-   11A RMS power calculation unit-   12 Band-pass filter-   13 High-pass filter-   14 Low-pass filter-   15 DC voltage detection unit-   16 DC voltage command generation unit-   16A DC voltage command generation unit-   16B DC voltage command generation unit-   16C DC voltage command generation unit-   17 DC voltage control unit-   18 Absolute value unit-   18 b Absolute value unit-   19 DC voltage value set table-   19B DC voltage value set table-   20 Phase calculation unit-   21 Three-phase to dq-axis conversion calculation unit-   22 a Multiplier-   22 b Multiplier-   23 Adder-   24 Subtracter-   25 Divider-   26 Limiter-   27 Multiplier-   28 Limiter-   29 Limiter-   30 Comparator-   31 Switching unit-   32 Correction gain calculation unit-   33 Multiplier

BEST MODE FOR CARRYING OUT THE INVENTION Embodiment 1

FIG. 1 is a block diagram showing an example of a configuration of anelectrical power conversion apparatus according to Embodiment 1 of thepresent invention. The electrical power conversion apparatus includes aconverter 2 that converts an AC power from a single-phase AC powersource 1 into a DC power, a capacitor 3 that stores the DC powerproduced by rectifying the AC power using the converter 2, an inverter 4that converts the DC power stored in the capacitor 3 into a three-phaseAC power of arbitrary frequency. The inverter 4 drives the inductionmachine 5 that is an AC rotating machine. The converter 2 controls ACpower from the AC power source 1 of the commercial frequency in a PWM(pulse width modulation) mode to convert it into the DC power. Theinverter 4 controls a low speed range of operation in avariable-voltage/variable-frequency (VVVF) mode and a high speed rangeof operation in a constant-voltage/variable-frequency (CVVF) mode.

Current detection units 6 a, 6 b and 6 c, which are current measuringinstruments on the AC side, detects phase currents iu, iv and iw thatflow in an induction machine 5, respectively. FIG. 1 describes thecurrent detection units 6 a, 6 b and 6 c on the AC side that detect bymeans of CT or the like the currents flowing through connection linesthat connect the inverter 4 to the induction machine 5; however, thephase currents may be detected through another known technique, usingcurrents flowing through the electrical power conversion apparatus, suchas bus currents. Since a relationship defined by an equation ofiu+iv+iw=0 holds and the w-phase current can be calculated from detectedcurrents for two other phases, u and v, a current detection unit 6 c forthe w-phase may be omitted.

A voltage control unit 7 determines the magnitude of AC voltagegenerated from the inverter 4 based on a torque current command valueIq, a magnetic flux current command value Id, and a rotation angularfrequency ω of the AC rotating machine. The angular frequency ω may bebased on speed information derived from a speed sensor mounted on theinduction machine 5. Alternatively, since there is a speed command valueω* available in a system in which the speed is controlled, the speedcommand value ω* may be used as the angular frequency ω. In addition,the angular frequency ω may be an estimated speed value that iscalculated in a speed sensorless control in which a speed sensor is notmounted.

FIG. 2 is a diagram illustrating a configuration of a ripple detectionunit 8 in the electrical power conversion apparatus according toEmbodiment 1 of the present invention. The ripple detection unit 8,which detects a ripple component that accompanies AC-DC conversion bythe converter 2, includes a root-mean-square (RMS) power calculationunit 11 and a band-pass filter 12. The RMS power calculation unit 11calculates a RMS power P that is to be generated from the inverter 4 bysumming together respective values calculated by multiplying Vu* by iu,Vv* by iv, and Vw* by iw, using phase currents iu, iv and iw that aredetected with the current detection unit 6 and voltage command valuesVu*, Vv*, Vw* that are calculated with the voltage control unit 7 andare to be generated from the inverter 4. The band-pass filter 12 filtersout a ripple component of the RMS power P generated from the RMS powercalculation unit 11. The RMS power calculation unit 11 calculates theRMS power P based on the following equation.

The RMS power, which is an output from RMS power calculation unit 11,includes a ripple component of a motor current resulting from the ripplecomponent that accompanies AC-DC conversion by the converter 2. Notethat the RMS power may be calculated using voltage and current values ina rotating orthogonal coordinate system.

A band-pass filter 12 in FIG. 2 derives only the ripple componentcontained in the RMS power P and accompanies AC-DC conversion by theconverter 2. When the AC power source 1 is a single-phase power sourceof the commercial frequency, the frequency of the single-phase powersource is 60 Hz or 50 Hz in Japan. Thus, the ripple component thataccompanies AC-DC conversion by the converter 2 is 120 Hz or 100 Hz infrequency, which is twice the frequency of the single-phase AC powersource.

In the present embodiment, the band-pass filter 12 is configuredassuming that the frequency of the single-phase AC power source is, forexample, 60 Hz. FIG. 3 shows a diagram illustrating the band-passfilter. The band-pass filter 12 is configured by a combination of ahigh-pass filter (HPF) 13 that passes therethrough frequencies higherthan those corresponding to a time constant T₁, which is a first timeconstant, and a low-pass filter (LPF) 14 that passes therethroughfrequencies higher than those corresponding to a time constant T₂, whichis a second time constant. The time constant T₁ of the high-pass filterand the time constant T₂ of the low-pass filter are determined such thatT₁=60 Hz and T₂=180 Hz in order to have a center frequency at 120 Hz.Namely, the time constants T₁ and T₂ are defined by Equations (2) and(3).

T ₁=1/(2π×60)   (2)

T ₂=1/(2π×180)   (3)

An example of a gain characteristic and a phase characteristic(generally called Bode diagram) in a frequency generated when theband-pass filter 12 of FIG. 3 is configured with the time constantsgiven by Equations (2) and (3), is as shown in FIG. 4. Thecharacteristics shown in FIG. 4 indicates that the gain characteristicis one that passes the frequencies around 120 Hz with almost noattenuation. For that reason, the band-pass filter 12 can derive the 120Hz component—a ripple component that accompanies AC-DC conversion by theconverter 2—to output a ripple component P_BEET.

Referring back to FIG. 1, the electrical power conversion apparatusaccording to the present invention includes a DC voltage commandgeneration unit 16 and a DC voltage control unit 17. The DC voltagecommand generation unit 16 is supplied with the rotation angularfrequency ω of the AC rotating machine, to generate a command value Vc*for a DC voltage Vc that is a voltage across the capacitor 3 to becharged by the converter 2 and that is measured by a DC voltagedetection unit 15 that is a voltage measuring instrument. The DC voltagecontrol unit 17 controls the converter 2 in accord with the commandvalue Vc*. The DC voltage command generation unit 16 increases the DCvoltage only during a time when the angular frequency ω generated fromthe inverter 4 has much influence, resulting from the ripple of the DCpower, on a torque and the like.

A mathematical analysis will be described below in terms of a reason inwhich the capacitance of the capacitor 3 can be reduced by controllingthe magnitude of the DC voltage according to the condition of theinverter 4 in accordance with the present invention.

Assuming that an input current iA to the converter 2 is a sinusoidalwave, an input power source voltage Vs and the input current iA to theconverter 2 are represented as follow:

V=√2 E cos(ωt+φ)   (4)

iA=√2 I cos(ωt)   (5)

From the above equations, an input power Pin to the converter 2 isrepresented by the following equation.

$\begin{matrix}\begin{matrix}{{{Pin} = {2\mspace{14mu} E\mspace{14mu} I\mspace{14mu} {\cos \left( {{\omega \; t} + \varphi} \right)}{\cos \left( {\omega \; t} \right)}}}\mspace{11mu}} \\{= {E\mspace{14mu} {I\left( {{\cos \left( {{2\omega \; t} + \varphi} \right)} + {\cos \; \varphi}} \right)}}}\end{matrix} & (6)\end{matrix}$

where constant terms in Equation (6) refers to a power supplied to aload and the sinusoidal wave component that varies with twice theangular frequency of ω refers to a ripple power supplied to thecapacitor 3. Since a power factor in the converter 2 is controlled to bea value of one, cos φ is 1.0, and the constant terms result in E×I.

Assuming that the ripple power component in Equation (6) is designatedPin˜, Pin˜ is represented by the following equation.

Pin˜=E I(cos(2ωt+φ))   (7)

On the other hand, the capacitance of the capacitor 3 is represented asC and the voltage across the capacitor 3, as Vc, and if it is assumedthat no influence due to the ripple of the voltage Vc across thecapacitor 3 occurs on the inverter 4 side, the following equation holds.Here, the voltage Vc across the capacitor 3 is called DC voltage.

$\begin{matrix}\left\lbrack {{Equation}{\mspace{11mu} \;}1} \right\rbrack & \; \\{{Pin}^{\sim} = {{\frac{}{t}\left( {\frac{1}{2} \cdot C \cdot {Vc}^{2}} \right)} = {{{Vc} \cdot C \cdot \frac{}{t}}{Vc}}}} & (8)\end{matrix}$

By substituting Equation (7) into Equation (8), the followingdifferential equation is expressed in terms of the DC voltage Vc.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack & \; \\{{{{Vc} \cdot C}\frac{}{t}{Vc}} = {E \cdot I \cdot {\cos \left( {{2\omega \; t} + \varphi} \right)}}} & (9)\end{matrix}$

Solving the differential equation of Equation (9), with an average valueVcav of the DC voltage as an initial value, provides a solution asbelow.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 3} \right\rbrack & \; \\\begin{matrix}{{Vc} = \sqrt{V_{cav}^{2} + {\frac{E \cdot I}{\omega \; C}{\sin \left( {{2\omega \; t} + \varphi} \right)}}}} \\{\cong {V_{cav} + {\frac{E \cdot I}{2\omega \; {CV}_{cav}}{\sin \left( {{2\omega \; t} + \varphi} \right)}}}}\end{matrix} & (10)\end{matrix}$

where Equation (10) assumes that a value of (E×I)/(2ωC×Vcav²) issufficiently smaller than a value of one, and an approximation is used,such that √(1+ε)≅1+ε/2 in terms of symbol ε that is sufficiently smallerthan a value of one.

The second terms of Equation (10) represents a ripple component of theDC voltage Vc. The ripple component represents a frequency of twice thepower source frequency, and it is seen that its magnitude is reverselyproportional to the capacitor capacitance C and the average value of theDC voltage Vc. The value (E×I) represents a power to be supplied to theconverter 2, and is maintained constant even though the DC voltage Vcvaries.

A current is flowing in the DC capacitor is obtained based on thefollowing equation.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 4} \right\rbrack & \; \\{i_{c} = {{C\frac{}{t}V_{c}} = {\frac{E \cdot I}{V_{cav}}{\cos \left( {{2\omega \; t} + \varphi} \right)}}}} & (11)\end{matrix}$

Calculating based on Equation (10) a ripple factor δ of the DC voltageused in Non-Patent Document 1, the ripple factor δ is as given inEquation (12).

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 5} \right\rbrack & \; \\\begin{matrix}{\delta = {\frac{\frac{E \cdot I}{2\omega \; {CV}_{cav}}}{V_{cav}} \times {100\mspace{11mu}\lbrack\%\rbrack}}} \\{= {\frac{E \cdot I}{2\omega \; {C\left( V_{cav} \right)}^{2}} \times {100\mspace{11mu}\lbrack\%\rbrack}}}\end{matrix} & (12)\end{matrix}$

Equation (12) shows that if a value of (E×I)/2ωC is constant and whenthe average value Vcav of the DC voltage Vc is increased, the ripplefactor can be reduced inversely proportionally to the square of thevalue Vcav. Further, Equation (12) indicates that when the ripple factoris assumed to remain the same, the capacitor capacitance C can bereduced if the average value Vcav of the DC voltage Vc is increased.Equation (12) shows that, by increasing the average value Vcav of the DCvoltage Vc by, for instance, 20% from 3000 V to 3600 V, the capacitorcapacitance, provided that the same ripple factor is applied, can bereduced approximately 30% (more specifically 30.6%).

Equations (10) and (12) are theoretical equations under assumption thatthe ripple of the DC voltage Vc has no influence on the output side ofthe inverter 4; however, they hold in a substantially similar fashioneven when the voltage generated from the inverter 4 contains a ripplecomponent.

Steady increase of the DC voltage results in an increase in a ratedvoltage of a switching device constituting the inverter 4, thus leadingto the use of a switching device having a higher voltage rating;therefore, there is a possibility of increasing costs. Further, evenwhen the switching device can be used which does not have a highervoltage rating, the use of the switching device with a higher voltagethan the rated voltage results in reduction of the lifetime of thedevice. In consideration of these factors, the DC voltage is increasedonly during times when the inverter 4 generates a voltage whose angularfrequency ω has much influence, resulting from the ripple of DC voltage,on the torque and the like.

When the inverter is operated in a one-pulse mode, the influence of theDC voltage on the switching device is reduced in comparison with amulti-pulse mode, even though the DC voltage is increased to more thanthe rated voltage, as will be described in detail later.

FIG. 5 is a diagram showing the DC current command generation unit inthe electrical power conversion apparatus according to Embodiment 1 ofthe present invention. The DC voltage command generation unit 16 isconfigured with an absolute value unit 18 that converts the angularfrequency ω to its absolute value, and a DC voltage value set table 19.The absolute value unit 18 takes an absolute value of the angularfrequency ω so that only a positive value is selected in order tosimplify the DC voltage value set table 19 because the angular frequencyω to be supplied is assigned a positive or negative sign. In the DCvoltage value set table 19, the angular frequency ω that turns theabsolute value is shown on the horizontal axis and a DC voltage commandvalue to be generated, on the vertical axis. In the DC voltage value settable 19 as shown in FIG. 15, the DC voltage is at its maximum voltageof 3600 V in a range (in the present embodiment, a range of 115 Hz to125 Hz, inclusive) including the frequency that is twice (in this case,the frequency is at 120 Hz; however, 100 Hz may in some cases be useddepending on an AC power source) the frequency of the AC power sourcewhere the voltage across the capacitor 3 contains a ripple component, inother words, there is present a large beating phenomenon. In theprevious range (a range of 60 Hz to 115 Hz, inclusive), the DC voltageis increased progressively, and in the subsequent range (a range of 125Hz to 180 Hz, inclusive), the DC voltage is decreased progressively.Progressively increasing and decreasing the DC voltage can reduce theburden of increasing the DC voltage on the switching device constitutingthe inverter 4. In the present embodiment, the DC voltage is increasedin the predetermined range (a range of 60 Hz to 180 Hz, inclusive). Therange where the DC voltage becomes a maximum is from 115 Hz to 125 Hz,inclusive.

In the DC voltage value set table 19, the predetermined range in whichthe DC voltage is increased more than usual is determined to be within arange where a beat rate β is permissible. The beat rate is based on thefollowing equation that is defined in Non-Patent Document 1.

β=(b−a)/a   (A)

where numeral b is a fluctuation range of an inverter output current,and numeral a is a fluctuation range of an output frequency of theinverter output current.

The upper limit value of a predetermined frequency range where the DCvoltage is increased more than usual needs to be a value where a beatrate at a normal DC voltage is permissible. Ditto for the lower limitvalue of the predetermined frequency range. In the frequency range wherethe DC voltage is increased more than usual, the range needed to beselected such that a beat rate of DC voltage is permissible at afrequency in the range. Extending the frequency range where the DCvoltage is increased more than usual ensures that the beat rate will bewithin the permissible range. The predetermined range may be determinedbased on another indicator other than the beat rate. It also may bedetermined in whatever manner that allows a ripple component of the RMSpower generated from an inverter to be controlled within the allowablerange.

The frequency range where the DC voltage is increased more than usual issuitably determined in consideration of values such as the beat rate βthat is allowable, a target value of the ripple factor δ at thefrequency of twice the AC power source frequency, and a ratio of anormal value over a maximum value of the DC voltage. Like the presentembodiment, when the beat rate β is 1.2 or less, the ripple factor atthe frequency of twice the AC power source frequency is 10%, and theratio of a normal value over a maximum value of the DC voltage is 1.2,it generally suffices if a width of the frequency range for maximizingthe value of the DC voltage is determined 10 Hz, as describedpreviously.

In the DC voltage value set table 19, the table data are configured suchthat the DC voltage command value Vc* does not exceed an over-voltageset value of the inverter 4. The maximum value of increasing the DCvoltage, 3600 V is determined in consideration of a rated voltage andcharacteristic of a switching device constituting the inverter 4.

The DC voltage control unit 17 receives the DC voltage command value Vc*that is an output from the DC voltage command generation unit 16 and theDC voltage Vc detected by the DC voltage detection unit 15. The controlunit 17 calculates the difference between the DC voltage command valueVc* and the DC voltage Vc, to control the converter 2 so as to producezero voltage difference between them.

Operation of the voltage control unit 7 that controls the voltagegenerated from the inverter 4 will be described. First of all, motorconstants of the induction machine, which are used for describing theoperation of the voltage control unit 7, are defined as below.

Rs Primary resistance value of motor

Ls Primary inductance of motor

M Mutual inductance of motor

Lr Secondary inductance of motor

Rr Secondary resistance value of motor

σ=1−M M/Ls/Lr

The voltage control unit 7 calculates a slip angular frequency commandvalue ωs* using the torque current command value Iq* and the magneticflux current command value Id* based on Equation (13).

ωs*=(Iq*/Id*)(Rr/Lr)   (13)

The inverter 4 calculates an inverter angular frequency ωinv thatcorresponds to the frequency of the voltage generated from the inverter4, by subtracting from a sum of the slip angular frequency command valueωs* and the frequency ω an amount of correction F_BEET obtained bymultiplying a predetermined coefficient Kf by a ripple amount P_BEETcalculated with the ripple detection unit 8. Namely, the inverterangular frequency ωinv is calculated based on Equation 14.

ωinv=ω+ωs*−F_BEET   (14)

F_BEET=Kf P_BEET   (15)

In this way, a frequency of the voltage generated from the inverter 4based on a ripple component derived from the ripple detection unit 8 iscorrected in the present embodiment 1. A d-axis voltage command valueVd* and a q-axis voltage command value Vq* on two rotational axes can becalculated using the inverter angular frequency ωinv, the torque currentcommand value Iq* and the magnetic flux current command value Id*, basedon Equation (16) and Equation (17).

Vd*=Rs Id*−ωinv σ Ls Iq*   (16)

Vq*=Rs Iq*+ωinv Ls Id*   (17)

As known by those skilled in the art, when the three-phase voltage orthree-phase current is coordinate converted into that of two orthogonalaxes of rotation, a coordinate control axis is needed. A phase of thecoordinate control axis—a rotating two-axis coordinate system based onthe angular frequency ω—is assigned θ. The phase θ is calculated basedon Equation (18) by integrating the inverter angular frequency ωinv.

[Equation 6]

θ=∫ωinv·dt   (18)

Since a voltage phase θv of the voltage command value is slightlyadvanced with respect to the phase θ, the voltage phase θv is calculatedbased on Equation (19).

θv=θ+tan⁻¹(Vd*/Vq*)   (19)

Three-phase voltage command values Vu*,Vv* and Vw* are calculated basedon Equation (20) using the voltage phase θv calculated using Equation(19), the d-axis voltage command value Vd* and the q-axis voltagecommand value Vq*.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 7} \right\rbrack & \; \\{\begin{pmatrix}{Vu}^{*} \\{Vv}^{*} \\{Vw}^{*}\end{pmatrix} = {\sqrt{\left( {Vd}^{*} \right)^{2} + \left( {Vq}^{*} \right)^{2}}\begin{pmatrix}{\cos \left( {\theta \; v} \right)} \\{\cos \left( {{\theta \; v} - {\frac{2}{3}\pi}} \right)} \\{\cos \left( {{\theta \; v} + {\frac{2}{3}\pi}} \right)}\end{pmatrix}}} & (20)\end{matrix}$

The inverter 4 performs DC-AC conversion based on the three-phasevoltage command values Vu*, Vv* and Vw*, calculated based on Equation(20) and obtained from the voltage control unit 7.

The frequency of voltage generated from the inverter 4 based on theripple component derived from the ripple detection unit 8 is corrected,thus enabling reductions of the motor current ripple on the output sideof the inverter 4 and associated torque ripple. FIG. 6 shows a graphillustrating an operation of the ripple detection unit and the DCcurrent command generation unit. It is assumed in FIG. 6 that Kf=1. Ifthe ripple component P_BEET, derived from the ripple detection unit 8 insynchronization with the ripple of the motor current on the output sideof the inverter 4 and associated torque ripple is positive in sign, thenadjustments are made so as to reduce the inverter angular frequencyωinv, and the frequency of the three-phase voltage command values Vu*,Vv* and Vw* that are generated from the voltage control unit 7 arereduced. Conversely, if the ripple component P_BEET, derived from theripple detection unit 8, is negative in sign, adjustments are made so asto increase the inverter angular frequency ωinv, and the frequencies ofthe three-phase voltage command values Vu*,Vv* and Vw* that aregenerated from the voltage control unit 7 are increased. This allows forthe control operation in response to the ripple of the motor current onthe output side of the inverter 4 and associated torque ripple, therebyreducing the motor current ripple and the torque ripple.

FIG. 7 is a set of graphs showing an effect of reducing a torque rippleby the electrical power conversion apparatus according to Embodiment 1of the present invention. FIG. 7( a) shows a torque waveform generatedby implementing Embodiment 1, FIG. 7( b) showing a torque waveformgenerated when the reduction of the torque ripple is not controlled. Thetorque waveforms shown in FIG. 7 are generated by simulations in whichthe DC voltage is set at 3600 V and the inverter frequency at 115 Hz. InFIG. 7( b) where the reduction of the torque ripple is not controlled,the torque waveform ripples at 120 Hz, which is twice the single-phasepower sour frequency, while in FIG. 7( a) where Embodiment 1 isimplemented, it can be recognized that the torque waveform has littleripple.

From the above description, by implementing Embodiment 1 the influenceof the ripple that accompanies the AC-DC conversion by the converter isdetected as the ripple component contained in the RMS power of theinverter, thus correcting the frequency of the voltage generated fromthe inverter, whereby an advantageous effect is provided in that thetorque ripple and the like is reduced.

Further, by increasing the DC voltage during a time when the invertergenerates the output frequency in which the beating phenomenon becomeslarge, that is, only during a time when the voltage across the capacitorcontains a ripple component, the capacitance of the capacitor can belowered which is necessary for reducing the beating phenomenon to withinthe permissible range. Consequently, compactness and cost reduction ofelectrical power conversion apparatus can be achieved.

In addition, this is also true for other embodiments.

Embodiment 2

FIG. 8 is a block diagram showing an example of a configuration of anelectrical power conversion apparatus according to Embodiment 2 of thepresent invention. FIG. 9 is a diagram illustrating a configuration of aripple detection unit in the electrical power conversion apparatusaccording to Embodiment 2 of the present invention. In Embodiment 2, aripple detection unit 8A, a voltage control unit 7A and a DC voltagecommand generation unit 16A are different compared to those inEmbodiment 1. In Embodiment 1, a RMS power is calculated from respectivethree-phase command values and respective three-phase currents, and aripple component is derived from the RMS power, to correct a frequencybased on the ripple component. In Embodiment 2, the ripple detectionunit 8A calculates a RMS power from dq-axis voltage command values anddq-axis currents, and the voltage control unit 7A corrects an amplitudeof the current command value according to the ripple component of theRMS power. The DC voltage command generation unit 16A operates so as tocontrol the DC voltage according to the calculated value of the RMSpower, and adjust the DC voltage to its normal value when the RMS powerP is small in quantity and within a range where its beating phenomenonis permissible. Note that configurations other than those described aresimilar to those in Embodiment 1 and pertinent figures use the samereference numerals as well. Only the differences will be describedherein.

In Embodiment 2, as shown in FIG. 9, the ripple detection unit 8A thatdetects the ripple component that accompanies AC-DC conversion by theconverter 2 includes a phase calculation unit 20, a three-phase todq-axis conversion calculation unit 21 and a RMS power calculation unit11A. The phase calculation unit 20 receives the angular frequency ω asan input, to calculate the phase θ by integrating, as shown in Equation(18), ωinv that is to be calculated as will be described later. Thethree-phase to dq axis conversion calculation unit 21 calculates d-qaxis currents Id and Iq by means of the phase currents iu, iv and iwdetected by the current detection unit 6, using the phase θ.

The RMS power calculation unit 11A calculates the RMS power P using thedq-axis currents Id and Iq calculated by the three-phase to dq-axisconversion calculation unit 21 and the dq-axis voltage command valuesVd* and Vq* calculated by the voltage control unit 7A, based on thefollowing equation.

P=Vd*Id+Vq*Iq   (21)

To calculate the values based on Equation (21), the RMS powercalculation unit 11A includes multipliers 22 a and 22 b, and an adder23, in which a value obtained by multiplication of Vd* and Id using themultiplier 22 a and a value obtained by multiplication of Vq* and Iqusing the multiplier 22 b are summed together using the adder 23 tothereby generate the output of the adder 23 as the RMS power P.

The RMS power P, which is an output of the RMS power calculation unit11A, contains a motor current ripple and a torque ripple componentresulting from the ripple component that accompanies the AC-DCconversion by the converter 2.

The RMS power P calculated using the RMS power calculation unit 11A isinput to the band-pass filter 12, and the output P_BEET from theband-pass filter is input to the voltage control unit.

The subtractor 24 subtracts the output P_BEET of the band-pass filterfrom the output of the RMS power calculation unit 11A, to output to theDC voltage command generation unit 16A the subtraction result as a RMSpower P that does not contains the ripple component.

The voltage control unit 7A calculates the slip angular frequencycommand value ωs* from the torque current command value Iq* and themagnetic flux current command value Id*, using the motor constant of theinduction machine. Namely, the slip angular frequency command value cos*is calculated using Equation (13) as with Embodiment 1.

By summing together the slip angular frequency command value ωs* and theangular frequency ω, the inverter 4 calculates the inverter angularfrequency ωinv that corresponds to a frequency of the voltage to begenerated. That is, the inverter angular frequency ωinv is calculatedbased on Equation (22) shown below.

ωinv=ω+ωs*   (22)

The d-axis voltage command value Vd* and the q-axis voltage commandvalue Vq* on the two rotating axes can be calculated from the inverterangular frequency ω the torque current command value Iq* and themagnetic flux current command value Id*. Namely, the d-axis voltagecommand value Vd* and the q-axis voltage command value Vq* arecalculated based on Equation (16) and Equation (17) as withEmbodiment 1. Since the voltage phase θv of the voltage command value isslightly advanced with respect to the phase θ, it is calculated based onEquation (19) as with Embodiment 1.

The three-phase voltage command values Vu*, Vv* and Vw* are calculatedbased on Equation (21) from the voltage phase θv obtained using Equation(19), the d-axis voltage command value Vd* and the q-axis voltagecommand value Vq*. The amplitudes of the three-phase voltage commandvalues are reduced by a correction amount V_BEET obtained by multiplyingthe ripple component P_BEET of the RMS power by a coefficient Kv.

$\begin{matrix}\left\lbrack {{Equation}\mspace{14mu} 8} \right\rbrack & \; \\{\begin{pmatrix}{Vu}^{*} \\{Vv}^{*} \\{Vw}^{*}\end{pmatrix} = {\sqrt{\left( {Vd}^{*} \right)^{2} + \left( {Vq}^{*} \right)^{2} - {V\_ BEET}}\begin{pmatrix}{\cos \left( {\theta \; v} \right)} \\{\cos \left( {{\theta \; v} - {\frac{2}{3}\pi}} \right)} \\{\cos \left( {{\theta \; v} + {\frac{2}{3}\pi}} \right)}\end{pmatrix}}} & (23) \\{{V\_ BEET} = {{Kv}\mspace{14mu} {P\_ BEET}}} & (24)\end{matrix}$

According to Equation (23), if the ripple component P_BEET, derived fromthe ripple detection unit 8A in synchronization with the motor currentand the torque ripple on the output side of the inverter 4, is positivein sign, then the amplitudes of the three-phase voltage command valuesVu*, Vv* and Vw* that are generated from the voltage control unit 7A arereduced. Conversely, if the ripple component P_BEET, derived from theripple detection unit 8A is negative in sign, then the amplitudes of thethree-phase voltage command values Vu*, Vv* and Vw* that are generatedfrom the voltage control unit 7A are increased, thereby enablingreductions of the motor current ripple on the output side of theinverter 4 and the associated torque ripple.

Note that since, in the present embodiment, the amplitudes of thevoltage command values are corrected, the DC voltage cannot be increasedto its maximum value even in the frequency zone that is in theconstant-voltage and variable-frequency (CVVF) control mode, and thus,the DC voltage needs to be lowered from the maximum value by a controlamount required for reducing the ripple.

The DC voltage command generation unit 16A receives the RMS power P,which is ripple-free and the output from the ripple detection unit 8A,and the angular frequency ω. The absolute value unit 18 and the DCvoltage value set table 19 are the same as those in Embodiment 1. Thepurpose of the DC voltage command generation unit 16A according toEmbodiment 2 is to reduce the burden on the switching deviceconstituting the inverter 4 by varying an amplitude for increasing thevoltage according to the RMS voltage, in addition to limiting a periodof time of increasing the DC voltage, more than that in Embodiment 1.The present embodiment is based on the fact that the beating phenomenonvaries depending on a power or torque generated by the motor, that is,the greater the power at the same speed, the larger the beatingphenomenon becomes. Conversely, when the power is small, its ripplefactor is within the allowable range even if the DC voltage is at therated voltage. FIG. 7 of Non-Patent Document 1 also shows that when thevoltage is constant, the larger the output from the converter, thelarger the ripple factor becomes.

An absolute value unit 18 b of the DC voltage command generation unit16A receives as an input the RMS power P having the positive or negativesigns, to take the absolute value of the RMS power P and then togenerate the power P as a RMS power value P1. A divider 25 divides thevalue P1 by a predetermined value (for instance, a maximum power) tooutput the coefficient Kp. The coefficient Kp is definitely limited towithin the range 0≅Kp≅1 by a limiter 26. A multiplier 27 performsmultiplication of an output value of the limiter 26 by an output valueof the DC voltage value set table 19, and the DC voltage command valueVc* thereby results in a value considered for the RMS power. Becausewhen the value calculated by the multiplier 27 is a small value such aszero, a small value such as 0 V is produced, a limiter 28 performs thevoltage limiting function so that the DC voltage command value Vc* fallsinto the range of 3000 V to 3600 V, inclusive.

Since, according to Equation (12), the ripple factor is proportional tothe RMS power output by the inverter and reversely proportional to theDC voltage squared, and if the coefficient Kp is made proportional tothe square root of the RMS power, the ripple factors are substantiallythe same regardless of the magnitude of the RMS power so long as the RMSpower is large.

In the present embodiment, the DC voltage command value increases as theRMS power increases; however, it is expected that a similar advantageouseffect is provided also by increasing the DC voltage command value whena value(s) other than the RMS power—such as the current value, thecommand torque value, the torque current command value, or the torquecurrent value—increase(s). This is true for embodiments below.

From the above description, by implementing Embodiment 2, theadvantageous effect is provided in that an influence resulting from theripple that accompanies the AC-DC conversion by the converter isdetected as a ripple component contained in the RMS power, thuscorrecting the amplitude of the voltage generated from the inverter,whereby the torque ripple and the like are reduced. Further, when theRMS power is smaller than the predetermined value, the DC voltage ismade to be a normal value, and when the RMS power is greater than thepredetermined value and if the RMS power is large, the DC voltage valueis made to increase, thereby providing an advantageous effect in thatthe burden on the switching device constituting the inverter can bereduced in addition to reduction of the capacitor capacitance.Consequently, another advantageous effect is provided in thatcompactness and cost reduction of an electrical power conversionapparatus is achieved.

Embodiment 3

FIG. 11 is a block diagram showing an example of a configuration of anelectrical power conversion apparatus according to Embodiment 3.Embodiment 3 is different only in a DC voltage command generation unit16B from Embodiment 2. In Embodiment 3, conditions where the DC voltageis controlled according to a value of the calculated RMS power isfurther limited to when the RMS power has a positive value, and then theDC voltage value set table 19 is executed. In other words, increasingthe DC voltage is limited to only the time during power operation, andduring regeneration operation the DC voltage is fixed at a rated voltageof 3000 V. The purpose of present embodiment is to reduce the beatingphenomenon to small size during regeneration operation in comparisonwith power operation, and to achieve more energy saving by transferringas much energy as possible back to the AC power source, from theviewpoint of energy conservation during regeneration operation. Notethat configurations other than those described are similar to those inEmbodiment 2 and pertinent figures use the same reference numerals aswell. Only the differences will be described herein.

FIG. 12 is a diagram showing a DC current command generation unit in theelectrical power conversion apparatus according to Embodiment 3 of thepresent invention. In comparison with FIG. 10 in Embodiment 2, added area limiter 29, a comparator 30, and a switching unit 31.

If the RMS power P is greater than zero, that is, during poweroperation, the comparator 30 generates an output signal of 1, so thatthe switching unit 31 is set to a position A. When the RMS power P iszero or less, that is, during coasting or regeneration operation, thecomparator 30 further generates an output signal of zero, so that theswitching unit 31 is set to a position B.

In order to fix the DC voltage at a rated value of 3000 V duringregeneration operation, the limiter 29, which is connected to a contactof the position B of the switching unit 31, performs the voltagelimiting function to prevent the DC voltage from becoming more than orless than 3000 V.

The signal for switching between settings A and B of the switching unit31 gains the same advantageous effect for not only the RMS power, butalso the torque command, power operation command, or regenerationoperation (braking) command.

In the present embodiment, the DC voltage command generation unit 16Bincludes the comparator 30, the limiter 29 and the switching unit 31, sothat increasing the DC voltage is limited only to the time during poweroperation and the DC voltage during regeneration operation is fixed at arated voltage of 3000 V. This limits the condition for increasing the DCvoltage, thereby providing an advantageous effect in that the burden onthe switching device constituting the inverter 4 can be reduced. Whenthe RMS power is small in amount, the DC voltage remains unchanged atthe normal value even during power operation, and even when the RMSpower is great in amount, the magnitude of increasing the DC voltage iscaused to vary according to the RMS power; however, even by increasingthe voltage during power operation independently of the magnitude of theRMS power, a similar advantageous effect is achieved if the DC voltageis not caused to increase.

Embodiment 4

FIG. 13 is a block diagram showing an example of a configuration of anelectrical power conversion apparatus according to Embodiment 4. FIG. 14is a diagram showing a ripple detection unit in the electrical powerconversion apparatus according to Embodiment 4 of the present invention.The differences in Embodiment 4 when compared to Embodiment 2 are aripple detection unit 8B and a DC voltage command generation unit 16C.

As with Embodiment 2, the ripple detection unit 8B includes thethree-phase to dq-axis conversion unit 21, the phase calculation unit20, the RMS power calculation unit 11A and the band-pass filter 12, andadditionally includes a correction gain calculation unit 32 thatreceives the angular frequency ω as an input, to calculate a correctiongain k, and a multiplier 33 that multiplies the correction gain k, whichis an output from the correction gain calculation unit 32, by an outputvalue of the band-pass filter 12.

The correction gain k is caused to vary with the angular frequency ω andthen may be determined using the table data, or may be provided in theform of a mathematical function. In particular, the correction gain isdetermined to become a maximum before the ripple frequency component of120 Hz, for instance. Further, if the correction gain is determinedzero, then no correction will be made, which provides an advantageouseffect in that, by varying the correction gain with respect to theangular frequency ω, it is determined whether or not a correction is tobe made, or if actually made, how much of the correction is required canbe varied according to the angular frequency ω.

FIG. 15 is a diagram showing the DC voltage unit command generation 16Cin the electrical power conversion apparatus according to Embodiment 4of the present invention. The ripple detection unit 16C is configuredwith an absolute value unit 18 that converts the angular frequency ω toits absolute value, and a DC voltage value set table 19B. Because theangular frequency ω to be supplied is assigned a positive or negativesign, the absolute value unit 18 converts the angular frequency ω to theabsolute value so that only a positive value is selected in order tosimplify the DC voltage value set table 19B. In the figure, the angularfrequency ω that has turned the absolute value is shown on thehorizontal axis and a DC voltage command value to be generated, on thevertical axis. In the DC voltage value set table 19B, the angularfrequency ω that has turned the absolute value is shown on thehorizontal axis and a DC voltage command value to be generated, on thevertical axis. In the DC voltage value set table 19B as shown in FIG.15, the DC voltage is at its maximum voltage of 3300 V in apredetermined range (a range of 115 Hz or more in the presentembodiment) including the frequency that is twice (in this case, thefrequency is at 120 Hz; however, 100 Hz may in some cases be useddepending on an AC power source) the AC power source frequency in whichthere is present a large beating phenomenon; and in the previous range,the DC voltage is increased progressively. An advantageous effect isthat keeping the voltage high also in a range of 120 Hz or moreeliminates operation of reducing the DC voltage, and anotheradvantageous effect is that a loss in a high speed range at 120 Hz ormore can be reduced because increasing the voltage reduces the motorcurrent flowing through the motor.

The output voltage of the inverter 4, that is, a motor voltage Vm iscontrolled so that a value of Vm/ω is substantially constant, in a rangewhere the frequency ω of the inverter 4 is above zero and under thepredetermined angular frequency (ω1) (i.e., a range of 0<ω<ω1). Afterthe motor voltage Vm has reached a maximum value shown based on thefollowing equation, the inverter 4 can no longer control an outputvoltage amplitude. The angular frequency at which the voltage reachesthe maximum value is ω1.

The value of ω1 is normally smaller than a frequency corresponding totwice the AC power source frequency. In a range where the angularfrequency ω is larger than ω1, the motor voltage Vm is fixed at itsmaximum value, and only the frequency varies.

Vm=√6/πVc   (25)

The characteristic of a maximum torque Tmax of the induction machinethat is controlled with this pattern of the motor voltage Vm is shown bythe following relationship in the high speed (ω>ω1) range.

Tmax∝(Vc/ω)²   (26)

When the DC voltage Vc is assumed constant, the maximum torque Tmax isreversely proportional to the square of the angular frequency ω. Thus,the torque is significantly reduced in, particularly, the high speedrange, so that the sufficient torque is difficult to obtain in the highspeed range.

Even without increasing a withstand voltage of the switching deviceconstituting the inverter 4, the DC voltage Vc can be increased in thehigh speed range (ω>ω1), as will be described next.

The switching device constituting the inverter 4 employs an insulatedgate bipolar transistor (IGBT) having a turn-on and off ability.

A peak value Vp of a collector-emitter voltage waveform Vce of the IGBTdevice, shown when the device shown in FIG. 16 interrupts a current I isrepresented empirically by the following equation.

Vp=Vc+I√(L/C)   (27)

where L is an inductive value of IGBT stray inductance, and C is a straycapacitance of IGBT stray capacitor.

In terms of a current value interrupted by the IGBT in an actualoperation, the one pulse mode is small compared to a multi-pulse(asynchronous) mode. The IGBT interrupts the current at its ripple peakvalue. Although the maximum value of current, Ip to be interrupted bythe IGBT in the multi-pulse (asynchronous) mode slightly variesdepending on factors such as the motor constant of induction machine, amodulation index of the output voltage generated from the inverter 4,and a length of wiring between the inverter 4 and the induction machine,the current value Ip, if a fundamental wave RMS value of the motorcurrent flowing through a motor is designated Im, is representedempirically as follows:

Ip=1.5×√2 Im   (28)

where the above coefficient 1.5 is normally a value on the order of 1.3to 1.5, and the upper limit value of 1.5 is used here.

On the other hand, in a motor waveform in the one pulse mode, aninterrupting current Iq corresponding to when the IGBT device interruptsthe current once per cycle is represented empirically as shown below, ifthe fundamental wave RMS value of the motor current is designated Im.

Iq=0.7×√2 Im   (29)

In Equation (28) and Equation (29), if no variation of the value Im isassumed to occur in any pulse mode, the following equation hold:

Ip≅2.1 Iq   (30)

When the pulse mode is transferred to the one pulse mode, the amount ofovercharging I×√(L/C) in Equation 27 becomes smaller and even if the DCvoltage Vc is increased, the peak value Vp of the IGBT collector-emittervoltage does not increase accordingly.

The inductive value L of the stray inductance of IGBT is approximately3.0 μH, and the stray capacitance C of the stray capacitor is in therange of approximately 1.5 μH to 3.0 μH. Assuming L=3.0 μH and C=1.5 μF,√(L/C) in Equation (27) results in a value of √2≅1.41. Even if the DCvoltage Vc is determined 3300 V as long as the current I interrupted bythe IGBT is 200 A or less, the Vp calculated based on Equation (27) willnot exceed 3600 V (the maximum value of voltage applicable to the IGBT).In particular, if the Ip is on the order of 100 A, the Vp results in theorder of 3450 V, thereby reducing an influence on the IGBT device. Thevoltages 3600 V and 3300 V are examples and those voltages aredetermined with considerations of a characteristics and serviceconditions of a switching device.

From Equation (30), the value Ip is approximately 257 A when Iq=120 A.From Equation (27), assuming Vc=3300 V and if I=120 A, the value Vp isapproximately 3470 A, and if I=257 A, the value Vp is approximately 3660V. In other words, in the case of the high speed range in the one pulsemode, the peak value Vp of IGBT collector-emitter voltage can be reducedto 3600 V or less; however, in the multi-pulse mode, there exists arange of the fundamental wave RMS value Im of the motor current, suchthat the value Vp exceeds 3600 V. If the value Im is in such range, theDC voltage in the low speed range is set to its normal value, and evenwhen the voltage is increased in a range above the speed range where thebeating phenomenon occurs, the withstand voltage of the switching deviceconstituting the inverter 4 does not need to be increased if the DCvoltage Vc is increased. Namely, when the DC voltage increases in a fullspeed range, the withstand voltage of the switching device needs to beincreased; however, even if the DC voltage is increased in a range abovethe speed range where the beating phenomenon occurs with the value inthe low speed range set to its normal value, the withstand voltage ofthe switching device does not need to be increased, thus achieving theinvention with the manufacture cost of the inverter 4 remaining thesame.

From the above description, by implementing Embodiment 4 the influenceof the ripple that accompanies AC-DC conversion by the converter isdetected as the ripple component contained in the RMS power of theinverter, thus correcting the amplitude of the voltage generated fromthe inverter, whereby an advantageous effect is provided in that torqueripples and the like are reduced.

Moreover, by increasing the DC voltage during the time when the invertergenerates the output frequency in which the beating phenomenon becomeslarge, the capacitance of the capacitor can be lowered which isnecessary for reducing the beating phenomenon to within the permissiblerange. Consequently, compactness and cost reduction of the electricalpower conversion apparatus is achieved. Further, by increasing the DCvoltage Vc in a frequency operated in the one pulse mode, the loss canbe reduced, and a greater torque can thereby be generated, withoutplacing an increased burden on the switching device.

The maximum value in the command value range of the DC voltage Vc isassumed 3300 V here; however, in the case of the 60 Hz power source inthe predetermined range including a frequency of a ripple component ofthe DC voltage Vc, the maximum value may be increased progressivelyfrom, for example, 85 Hz, and set at 3600 V from 115 Hz to 125 Hz, whileit may be reduced progressively at a frequency above 125 Hz, and set at3300 V for 140 Hz or more. In a frequency in which the inverter isoperated in the one pulse mode, the voltage is determined to be in arange where the command value of the DC voltage Vc is higher than normaland the burden on the switching device is not increased, whereby anadvantageous effect is provided in that the loss can be reduced and agreater torque can thereby be generated. Note that an upper limitvoltage value (3300 V, for instance) in the range where the commandvalue of the DC voltage Vc is higher than its normal voltage and theburden on the switching device is not increased, is called one-pulsemode upper limit voltage value. The one-pulse mode upper limit voltagevalue is determined so that a voltage to be applied to the switchingdevice under assumed service conditions does not exceed its maximumvalue. If, in a frequency in which the inverter is operated in the onepulse mode, the command value of the DC voltage Vc is determined to behigher than normal but lower than the one-pulse mode upper limit voltagevalue, the voltage value may vary with the variation in frequency, andthe command value may temporarily be the normal voltage.

It should be noted that in each embodiment above, the example of an ACrotating machine (induction machine) is shown as a load connected to theinverter 4; however, the AC rotating machine is not limited to aninduction machine. It is expected that a similar advantageous effectwill be achieved for another example where the invention is not limitedto an AC rotating machine and is applied to an apparatus that controlsanother load, for instance, an electromagnetic actuator such as a linearinduction motor, a linear synchronous motor, or a solenoid.

INDUSTRIAL APPLICABILITY

The present invention pertains to an inverter that variably drives an ACmotor using DC power, as an electrical source, obtained by rectifying anAC power source with a converter. In particular, the inverter is usedfor an electric railcar on a railway track of a single-phase AC powersource where ripple components generated by rectification increases. Theinvention is applicable to an apparatus that is a consumer productoperating on a single-phase power and controls a motor using aninverter, such as an air-conditioner, a refrigerator and a washingmachine.

1. An electrical power conversion apparatus comprising: a converter thatconverts an AC power from an AC power source into a DC power; acapacitor that stores the DC power produced from the converter; aninverter that converts to an AC power the DC power stored in thecapacitor; a voltage control unit that calculates command values for anAC voltage to be generated from the inverter, to control the inverter sothat the command values is produced; a current measuring instrument thatmeasures an AC current generated from the inverter; a ripple detectionunit that receives the AC voltage command values calculated by thevoltage control unit and the AC currents measured by the currentmeasuring instrument, to detect a ripple of an RMS power generated fromthe inverter; a voltage measuring instrument that measures the voltageacross the capacitor; a DC voltage command generation unit thatcalculates a command value of the voltage across the capacitor accordingto a frequency of the AC voltage generated from the inverter; and a DCvoltage control unit that receives the voltage measured by the voltagemeasuring instrument and the command value calculated by the DC voltagecommand generation unit, to control the converter so that the voltageacross the capacitor becomes the command value, wherein the DC voltagecommand generation unit makes the command values of the voltage acrossthe capacitor higher than usual in a situation where the frequency of ACvoltage generated from the inverter is within a predetermined rangeincluding a frequency of a ripple component of the voltage across thecapacitor, and the voltage control unit receives the ripple componentderived from the ripple detection unit, to calculate the command valuesfor the AC voltage generated from the inverter so that the ripplecomponent is reduced.
 2. The electrical power conversion apparatus asrecited in claim 1, wherein the predetermined range is determined to bethat in which the ripple component is permissible.
 3. The electricalpower conversion apparatus as recited in claim 1, wherein, within thepredetermined range, the command value is made higher than normal in atleast part of a frequency range in which the inverter operates in a onepulse mode, and made below an upper limit of a voltage value in a rangewhere a burden on a switching device of the inverter is not increased.4. The electrical power conversion apparatus as recited in claim 1,wherein the DC voltage command generation unit determines that thevoltage of the capacitor is a normal value in a situation where anabsolute value of the RMS power output by the inverter is below apredetermined value.
 5. The electrical power conversion apparatus asrecited in claim 1, wherein the DC voltage command generation unitdetermines that the voltage across the capacitor is a normal value in asituation where the RMS power output by the inverter is a negativevalue.
 6. The electrical power conversion apparatus as recited in claim1, wherein the DC voltage command generation unit includes a DC voltagevalue set table for calculating the command value based on the frequencyof the AC voltage generated from the inverter, the DC voltage value settable including a range for maximizing the command value in thepredetermined range, the command value increasing with an increase infrequency in a range lower than the range where the command value ismaximized within the predetermined range.
 7. The electrical powerconversion apparatus as recited in claim 6 wherein the command value isdecreased with an increase in frequency in a range lower than the rangewhere the command value is maximized within the predetermined range. 8.The electrical power conversion apparatus as recited in claim 1, whereinthe ripple detection unit includes a RMS power calculation unit thatcalculates a RMS power output by the inverter and a band-pass filterthat filters out a ripple from an output from the RMS power calculationunit.
 9. The electrical power conversion apparatus as recited in claim8, wherein the RMS power calculation unit calculates a RMS power bysumming together values obtained by multiplying respective ACthree-phase currents measured with the current measuring instrument byrespective AC three-phase voltage command values calculated by thevoltage control unit.
 10. The electrical power conversion apparatus asrecited in claim 8, wherein the RMS power calculation unit calculates aRMS power by summing together values obtained by multiplying respectivevalues in a rotating orthogonal coordinate system converted from thethree-phase AC currents measured with the current measuring instrument,by respective voltage command values in the rotating orthogonalcoordinate system calculated by the voltage control unit.
 11. Theelectrical power conversion apparatus as recited in claim 8, wherein theband-pass filter is configured by connecting in series a high-passfilter that includes a first first-order lag filter with a first timeconstant, for defining a lower limit of a passband frequency, and asubtractor that subtracts an output of the first-order lag filter froman input thereof, and a low-pass filter that includes a secondfirst-order filter with a second time constant, for defining an upperlimit of the passband frequency.
 12. The electrical power conversionapparatus as recited in claim 8, wherein the ripple detection unitincludes a correction gain calculation unit that calculates a correctiongain, and a multiplier that multiplies an output from the band-passfilter by the correction gain generated from the correction gaincalculation unit, and wherein an output from the multiplier is that fromthe ripple detection unit.
 13. The electrical power conversion apparatusas recited in claim 12, wherein the correction gain that is generatedfrom the correction gain calculation unit varies depending on thefrequency of the AC voltage generated from the inverter.
 14. Theelectrical power conversion apparatus as recited in claim 1, wherein thevoltage control unit controls, according to the ripple component,command values for the frequency of the AC voltage that is generatedfrom the inverter.
 15. The electrical power conversion apparatus asrecited in claim 1, wherein the voltage control unit controls, accordingto the ripple component, command values for an amplitude of the ACvoltage that is generated from the inverter according to the ripplecomponent.